TVRO receiver system with low-cost video-noise reduction filter

ABSTRACT

A TVRO receiver for receiving frequency-modulated video signals, the receiver comprising a tuner including a superheterodyne circuit having a voltage-controlled oscillator (VCO), means for supplying a controlling input voltage to the VCO, and a mixer for combining incoming 1st IF signals with the output of the VCO to reduce the frequency of the 1st IF signals to a 2nd IF frequency which is sufficiently high to permit the output frequency of the VCO to be above the frequency range of the 1st IF signals, and a linear phase passband filter for passing a single video channel in the 2nd IF output from the mixer, the filter producing an output which is at least about 10 dB down from its peak at both +10 MHz and -10 MHz from the center of the passband of the filter.

This is a continuation of co-pending application Ser. No. 792,767 filedon Oct. 30, 1985 now U.S. Pat. No. 4,711,117

BACKGROUND OF THE INVENTION

This invention relates generally to TVRO receivers for the reception ofa wide range of satellite TV signals and, more particularly, tofiltering means for reducing terrestrial interference (TI) and othervideo noise in a TVRO receiver system.

In a TVRO system, the satellite signals are received by an antenna(usually a paraboloidal dish) and converted to a lower "lst IF"frequencyat the antenna site. This conversion may be effected by a downconverter, which converts only a single channel to the 1st IF frequency,or a block converter, which converts all channels of a common polarityto a 1st IF block of frequencies ranging from 950 to 1450 MHz. Thisentire block of frequencies is then fed via coaxial cable to thereceiver, which selects a particular channel for viewing and/orlistening. In the receiver, the 1st IF signals are converted to a 2nd IFfrequency range which traditionally has been centered at 70 MHz in mostTVRO systems.

In the reception of TV signals broadcast by satellites, a major problemfaced by conventional TVRO receivers is distortion due to terrestrialinterference (TI) caused by the presence of local terrestrial microwavecommunication links. TI can lead to substantial degradation and eventotal loss of the signals received from a satellite. Previous approachesto this problem have involved the use of various filters within the TVROreceivers to perform wavetrapping within the interference region. Theuse of such filters is described, for example, in Battle et al., "How toIdentify and Eliminate Terrestrial Interference", TVRO Technology, May,1985, pp. 32-41.

The provision of TI filters is not a major challenge in TVRO receiversoperating at 70 MHz or similarly low 2nd IF frequencies. However, thepresently emerging generation of TVRO receivers use a higher 2nd IF,e.g., 612 MHz, so that the VCO frequencies are above the lst IF block offrequencies to prevent the VCO from interfering with the desiredsignals. At these higher 2nd IF frequencies, it is extremely difficultto produce trap or notch filters with the high Q required to eliminateTI and other video noise without excessive losses in the desired videosignal.

SUMMARY OF THE INVENTION

It is a primary object of the present invention to provide an improvedTVRO receiver which reduces the effects of terrestrial interference andother video noise in a TVRO system using a 2nd IF frequency high enoughto permit the 2nd IF VCO frequency to be above the lst IF frequencyrange.

A related object is to provide a TVRO receiver system with an improvedvideo-noise-reduction (VNR) filter, which can be manufactured at a lowcost.

Another object of this invention is to provide a TVRO receiver systemwhich rejects a significant amount of TI without requiring narrow notchor trapping filters at the TI center frequencies.

Other objects and advantages of the invention will be apparent from thefollowing detailed description and the accompaying drawings.

In accordance with the present invention, a TVRO receiver is providedwith a tuner including a superheterodyne circuit having avoltage-controlled oscillator (VCO), means for supplying a controllinginput voltage to the VCO, and a mixer for combining incoming lst IFsignals with the output of the VCO to reduce the frequency of the 1st IFsignals to a 2nd IF frequency which is sufficiently high to permit theoutput frequency of the VCO to be above the frequency range of the lstIF signals; and a linear phase pass band filter for passing a singlevideo channel in the 2nd IF output from the mixer, the filter producingan output which is at least about 10 db down from its peak at both +10MHz and -10 MHz from the center of the pass band of the filter.

In another aspect of the invention, the center of the signal spectrum ofthe 2nd IF video signal produced by the superheterodyne circuit in thetuner is offset from the 2nd IF center frequency, and the 2nd IF signalsare passed through a pass band filter which has the center of its passband aligned with the center of the signal spectrum of the 2nd IF videosignal.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention and further objects and advantages thereof may best beunderstood by reference to the following description taken inconjunction with the accompanying drawings, in which:

FIG. 1 is a simplified block diagram of a conventional TVRO earthstation;

FIG. 2 is a block diagram of a tuner for use in the TVRO system of FIG.1;

FIG. 3 is a block diagram of a demodulator for use in the TVRO system ofFIG. 1;

FIG. 4 is a diagrammatic illustration of exemplary signal spectra frommultiple satellite transponders having adjacent center frequencies, andfrom multiple terrestrial microwave communication channels having centerfrequencies located between those of the satellite channels;

FIG. 5 is an example of a satellite video signal spectrum with TI at oneedge thereof;

FIG. 6 is an example of the response characteristics one one exemplaryfilter used in accordance with the invention;

FIG. 7 is another example of a satellite video signal spectrum, with TIat both edges;

FIG. 8 is a schematic diagram of a preferred embodiment of anelectronically tunable VNR filter for use, in accordance with thepresent invention, in the TVRO system of FIG. 1;

FIG. 9 is a schematic diagram of a modified embodiment of the tunableVNR filter of FIG. 8;

FIG. 10 is a graphical representation of several exemplarycharacteristic responses of the filters shown in FIGS. 8 and 9;

FIG. 11 is a graphical representation of typical input vs. outputsignal-to-noise characteristics for the TVRO receiver system of FIG. 1;and

FIG. 12 is a schematic diagram of an electronically switchable VNRfilter for use, according to the system of the invention, with the TVROreceiver system of FIG. 1.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Although the invention will be described in connection with certainpreferred embodiments, it will be understood that it is not intended tolimit the invention to those particular embodiments. On the contrary, itis intended to cover all alternatives, modifications and equivalentarrangements as may be included within the spirit and scope of theinvention as defined by the appended claims.

Referring now to the drawings, in FIG. 1 there is shown a functionalblock diagram of a TVRO earth station for the reception of satellitesignals. The system includes an antenna 11, which is typically aparaboloidal dish equipped with a low noise block (LNB) converter andrelated accessories and positioning mechanisms, for capturing signalstransmitted from orbiting satellites; and a receiver system including atuner 12, a demodulator 13, a video processing and amplification section14, and an audio tuner 15.

The antenna 11 receives signals transmitted from the satellite in thefour-GHz frequency band (3.7 to 4.2 GHz); and this entire block offrequencies is converted to a 1st IF frequency range of 950 to 1450 MHzby the block converter located at the antenna site. The 1st IF signalsare then sent via coaxial cable to the tuner 12 which selects aparticular channel for viewing and converts the signals in thatparticular channel to a 2nd IF frequency range. The 2nd IF frequencyrange is preferably high enough to permit the 2nd IF VCO frequencies tobe above the 1st IF block of frequencies, to prevent the VCO frominterfering with the desired signals. For a lst IF frequency range of950 to 1450 MHz, this means that the center frequency of the second IFfrequency range must be at least 500 MHz. A particularly preferred 2ndIF center frequency in the system of the present invention is 612 MHz.

In the demodulator 13, the 2nd IF signal is passed through an amplifierand a filter and on to a conventional video detector which demodulatesthe frequency-modulated signal to the baseband of the original videosignal (e.g., 0 to 10 MHz), producing a composite video signal output.The filter preferably has a pass band that is only about 22 MHz wide; apass band of this width passes the essential video and audio informationwhile rejecting unwanted noise received by the antenna on the edges ofthe selected channel.

The output of the demodulator comprises the baseband signals which rangefrom DC to about 8.5 MHz; this includes video information from about 15KHz to 4.2 MHz, and subcarriers from about 4.5 to 8.5 MHz.

FIG. 2 shows a simplified block diagram of a suitable tuner 12 for usein the TVRO system of FIG. 1. This tuner 12 includes a passband filter19 having a passband that is 500 MHz wide (to pass signals in the 1st IFrange of 950 to 1450 MHz). From the filter 19, the lst IF signals arepassed through a preamplifier 20 to a superheterodyne circuit includinga voltage-controlled oscillator (VCO) 21 receiving a controlling inputvoltage on line 22, and a mixer 23 for combining the output of the VCO21 with the lst IF output of the amplifier 20. This converts of the 1stIF signals to a desired 2nd IF frequency range. The resulting 2nd IFsignals are passed through a pair of amplifiers 24 and 25 and then on tothe demodulator 13.

By adjusting the controlling input voltage supplied to the VCO 21 vialine 22, different channels (frequency bands) in the lst IF signals arecentered on the center frequency of the 2nd IF output of the mixer 23.Each channel typically contains at least a video carrier signal, a colorsubcarrier signal, and an audio signal at different prescribedfrequencies. These carrier and subcarrier signals for all the channelsare transmitted simultaneously from the satellite to the earth stationantenna 10 and block converter 11, and then over a cable to the tuner12.

The following "Table I" is a list of the center frequencies for 24transponders on a single satellite. Table I also lists the correspondingcenter frequencies in the output from the block converter 11 (identifiedin Table I as the lst IF center frequencies) and the output frequenciesrequired from the VCO 21 in order to tune the receiver to eachindividual transponder. It will be noted that the difference between the1st IF center frequency and the corresponding VCO output frequency foreach transponder is 612 MHz, which means that the center frequency ofthe 2nd IF output from the mixer 23 is 612 MHz for every transponder.That is, the VCO output frequencies listed in Table I will cause the612-MHz output frequency of the mixer 23 to be centered on thecorresponding 1st IF center frequency. For example, a VCO outputfrequency of 2042 MHz will cause the 612-MHz output frequency of themixer to be centered on the 1430-MHz lst IF center frequency oftransponder No. 1. A preferred system for controlling the input voltageto the VCO 21 to produce the desired output frequencies listed above isdescribed in Ma et al. copending U.S. Pat. application Ser. No.06/792,767 now U.S. Pat. No. 4,718,117 filed 10/85.

                  TABLE I                                                         ______________________________________                                        Transponder         1st IF    VCO     2nd IF                                  Number   Transponder                                                                              Center    Output  Center                                  ("Channel")                                                                            Center Freq.                                                                             Freq.     Freq.   Freq.                                   ______________________________________                                         1       3720 MHz   1430 MHz  2042 MHz                                                                              612 MHz                                  2       3740       1410      2022    612                                      3       3760       1390      2002    612                                      4       3780       1370      1982    612                                      5       3800       1350      1962    612                                      6       3820       1330      1942    612                                      7       3840       1310      1922    612                                      8       3860       1290      1902    612                                      9       3880       1270      1882    612                                     10       3900       1250      1862    612                                     11       3920       1230      1842    612                                     12       3940       1210      1822    612                                     13       3960       1190      1802    612                                     14       3980       1170      1782    612                                     15       4000       1150      1762    612                                     16       4020       1130      1742    612                                     17       4040       1110      1722    612                                     18       4060       1090      1702    612                                     19       4080       1070      1682    612                                     20       4100       1050      1662    612                                     21       4120       1030      1642    612                                     22       4140       1010      1622    612                                     23       4160        990      1602    612                                     24       4180        970      1582    612                                     ______________________________________                                    

FIG. 3 is a block diagram of a demodulator 13 for receiving the 2nd IFoutput of the tuner 12 in the TVRO system of FIG. 1. This demodulatorcircuit includes a pair of conventional IF amplifiers 30 and 31 forreceiving the 2nd IF signal from the final amplifier 25 in the tuner 12.Both of these amplifiers 30 and 31 receive an automatic gain control(AGC) signal from an input terminal 32. From the amplifier 31, the 2ndIF signal is passed through a filter 33 and on to a conventional videodetector 34 which demodulates the frequency-modulated signal to thebaseband of the original video signal (e.g , 0 to 10 MHz), producing acomposite video output signal. The 2nd IF filter 33 preferably has apass band that is only about 22 MHz wide; a pass band of this widthpasses the essential video and audio information while rejectingunwanted noise received by the antenna on the edges of the selectedchannel.

The AGC feedback loop includes an IF amplifier 36 which amplifies theoutput of the filter 33 and supplies it to an AGC detector 37. Theoutput of this detector 37 is passed through an AGC amplifier 38, whichproduces a signal strength meter drive signal at a terminal 39. Thissignal strength meter is usually located on the front panel of the TVROreceiver.

The illustrative demodulator also includes an IF amplifier 40 whichreceives the same input supplied to the video detector 34, amplifies it,and passes it through a narrow passband filter 41. The output of thefilter 41 is passed through a detector in the form of a diode 42. Thesignal passed by the diode 42 is smoothed by an amplifier 43 to producea DC output voltage that can be used to detect the presence of a signalnear the center frequency of the particular satellite channel to whichthe receiver is tuned. Although this signal is not used in the system ofthe present invention, it is useful to discriminate between satellitesignals and TI.

The output of the demodulator illustrated in FIG. 3 comprises thebaseband signals which range from DC to about 8.5 MHz; this includesvideo information from about 15 KHz to 4.2 MHz, and subcarriers fromabout 4.5 to 8.5 MHz. The video information in these baseband signals ispassed through the video processing and amplification section 14 beforebeing displayed on a video monitor or television set, and the audiosignals are passed through the audio tuner 15 and then on to one or morespeakers which convert the signals to audible sound.

In terrestrial video broadcasts, the video information is normally inthe form of amplitude-modulated ("AM") signals. In satellite videotransmissions, on the other hand, the video information is in the formof frequency-modulated ("FM") signals for line-of-sight transmissionover long distances. In FM video transmissions, the center (i.e., thepeak) of the signal spectrum is offset from the nominal center frequencyof the signal because of the synchronizing and blanking pulses thataccompany each segment of video information.

In accordance with one aspect of the present invention, the center ofthe pass band of the 2nd IF filter 33 is offset from the 2nd IF centerfrequency (e.g., 612 MHz) so that the center of the pass band is alignedwith the center of the 2nd IF signal spectrum. By centering the filterpass band on the video signal spectrum rather than the nominal centerfrequency of the frequency-modulated video signal, terrestrialinterference and other noise on one side of the signal spectrum isrejected by the filter. This significantly improves the quality of thefiltered video signal in those locations where TI or other noise islocated on the edge of the video signal spectrum that is closer to thenominal center frequency of the channel.

As illustrated in FIG. 4, the carrier frequencies allocated forsatellite and terrestrial communications are staggered in an effort tominimize interference. Thus, terrestrial carriers are odd multiples of10 MHz, while satellite carriers are even multiples of 10 MHz. On anygiven satellite, the signals from alternate transponders have a firstpolarization, and the signals from the intervning transponders have asecond polarization orthogonal to the first. As will be appreciated fromFIG. 4, TI will always be located 10 MHz away from the center frequencyof any given transponder. Because the center of the video signalspectrum in a satellite signal is usually offset from the centerfrequency by 1 to 3 MHz, TI will normally be either 7-9 MHz or 11-13 MHzaway from the center of the signal spectrum.

FIG. 5 is an actual example of a satellite video signal spectrum inwhich the center of the signal spectrum is offset from the transpondercenter frequency by about 3 MHz, and TI is present at the left edge ofthe spectrum, 10 MHz from the channel center frequency. It can be seenthat the strength of the TI signal is is about the same as that of thesatellite signal. The pass band of the filter 33 is indicated by thepair of broken lines in FIG. 5. It can be seen that this pass band,which is centered on the center of the signal spectrum and is 22 MHzwide, passes virtually all the video information in the satellite signalwhile rejecting the TI. The TI is rejected because it is a narrow bandsignal located about 13 MHz from the center of the signal spectrum,while the edge of the filter pass band is only 11 MHz from the center ofthe signal spectrum. It can also be seen from FIG. 5 that if the filterpass band were centered on the channel center frequency rather than thesignal spectrum, the TI would be passed through the filter along withthe video information.

In accordance with another aspect of this invention, the 2nd IF filteris a linear phase pass band filter having a pass band profile which isat least about 10 dB down at both +10 MHz and -10 MHz from the center ofthe pass band. This type of filter is commonly referred to as a"haystack" filter because the rounded shape of the filter responsecharacteristic (i.e., the pass band profile) resembles the profile of ahaystack. An actual example of the response characteristic of one suchfilter is shown in FIG. 6. Because of its sharply sloping band edges,the haystack filter rejects a major portion of the TI and other noise atboth edges of the video signal spectrum. This TI rejection is in sharpcontrast to the performance of the SAW filters which are commcnly usedin video demodulators; the SAW filters have generally square orrectangular response characteristics which pass substantial levels of TIeven at the very edges of the pass band.

FIG. 7 is another actual example of a satellite video signal spectrum inwhich the center of the signal spectrum is offset from the transpondercenter frequency by only about 1 MHz, and TI is present at both edges ofthe spectrum. It can be seen that the strength of each TI signal issubstantial, and in fact the TI closer to the center frequency isconsiderably stronger than the satellite signal. The responsecharacteristic of the haystack filter is such that it passes virtuallyall the video information in the satellite signal while rejecting majorportions of both the TI signals. It can be seen from FIG. 7 that a SAWfilter response characteristic would pass all the TI on the upper edgeof the video signal spectrum and a major portion of the TI signal on thelower edge of the spectrum.

FIG. 8 is a schematic diagram of an exemplary video-noise-reductionfilter 50 for use in accordance with the system of this invention. Thisfilter 50 essentially comprises a first notch section 51 and a secondnotch section 52 connected in parallel across a length of striptransmission line 53 and a reference ground plane 54. The striptransmission line 53 is one of the conventional 50-ohm type and has bothits ends connected to the reference ground plane.

The resonance of the resonant circuits within the two notch sections 51and 52 is electronically controlled by the use of voltage-controlledvariablereactance devices. More specifically, variable-reactance orvaractor diodes are used within each of the notch sections to controlthe respective notch frequencies by electrical means. Thus, the firstnotch section 51 consists of a series resonant circuit comprising aninductance 55, a capacitance 56, and a varactor diode 57. A d-c blockingcapacitance 58 is used within the notch section 51 to connect thecathode end of the varactor diode 57 within the resonant circuit to thereference ground plane at radio frequencies. The second notch section 52is similar to the first one and consists of a series resonant circuitcomprising an inductance 59, a capacitance 60 and a varactor diode 61.

A first tuning voltage V_(T1) is applied through a radio-frequency choke63 to the anode of the varactor diode 57. A tuning capacitor 64 alsoconnects the junction of the radio-frequency choke 63 and the anode ofthe varactor diode 57 to the reference ground plane.

A second tuning voltage V_(T2) is applied to the cathode of the varactordiode 57 within the notch section 51 through a series connection of ablocking capacitance 65 and a limiting resistance 66. The same tuningvoltage V_(T2) is also applied to the anode of the varactor diode 61within the notch section 52 through a series connection of a blockingcapacitor 67 and a limiting resistance 68. The cathode of the varactordiode 61 is also connected to the reference ground plane through aparallel connection of a radio-frequency choke 69 and a tuningcapacitance 69a.

The function and operation of varactor diodes is well known. A varactordiode makes use of the change in capacitance of a reverse-biased pnjunction as a function of applied voltage. The diode conducts normallyin the forward direction, but the reverse current saturates at arelatively low voltage and then remains constant, eventually risingrapidly at the avalanche point. In controlling the variation in resonantfrequency of the notch sections 41 and 42 of the filter 40, theoperating region of interest for the varactor diodes lies between thereverse saturation point, which gives the maximum junction capacitance,and a point just above avalanche, at which the minimum diode capacitanceis obtained. Thus, the limits on the range of capacitance obtained fromthe diode are prescribed by the two conditions limiting the reversevoltage swing, i.e., conduction and avalanche. Ideally, the varactordiodes to be used with the VNR filter, according to the preferredembodiment, should have a large capacitance variation and a small valueof minimum junction capacitance, in addition to the lowest possiblevalue of base resistance which leads to low noise.

Returning to the filter shown in FIG. 8, the two tuning voltages V_(T1)and V_(T2) are applied to the anode and cathode, respectively, of thevaractor diode 57 within the first notch section 51, and serve to definethe operating point of the varactor diode on its characteristic curve,by controlling the reverse-bias across the diode junction. Morespecifically, the first tuning voltage V_(T1) has a fixed value andreverse-biases the varactor diode 57 by a fixed amount. The diode 57 atthis point presents a certain value of junction capacitance into theresonant circuit of the first notch section 51. The value of the tuningvoltage V_(T1) is chosen in such a way that the resonant circuitresonates around a center frequency which corresponds to theintermediate frequency (IF) used by the filter.

Subsequently, the second tuning voltage V_(T2) is applied to thevaractor diode 57 and this voltage serves to change the amount ofreverse-bias effective across the diode junction, which in turn producesa change in the junction capacitance presented to the resonant circuitby the varactor diode 57. The second tuning voltage V_(T2) can thus bevaried to adjust the notch frequency of the first notch section 51.

The second tuning voltage V_(T2) is also applied to the varactor diode61 in the second notch section 52 of the filter 50 in such a way as toreverse-bias the diode junction. Any variation in the voltage V_(T2)produces a corresponding change in the junction capacitance of thevaractor diode 61 and hence the resonant frequency of the resonantcircuit of the second notch section 52. Thus the voltage V_(T2) can bevaried to adjust the notch frequency of the second notch section 52.

In effect, the second tuning voltage V_(T2) can be used to adjust thewidth and profile of the pass band of the filter with respect to itscenter frequency. Even more importantly, the notch frequencies of thetwo notch sections 51 and 52 can be adjusted in such a way that thefilter response characteristic is at least 10 dB down from its peak atboth +10 MHz and -10 MHz from the center of the desired signal spectrumin the pass band.

FIG. 9 is a schematic diagram of a possible variation of the tunable VNRfilter shown in FIG. 8. The embodiment of FIG. 9 differs only in thenumber and symmetrical arrangement of components from the embodiment ofFIG. 8.

The filter 70 of FIG. 9 essentially comprises a first notch section 71and a second notch section 72 connected in parallel fashion across alength of strip transmission line 73 and a reference ground plane 74. Asin FIG. 8, the strip transmission line 73 is of the conventional 50-ohmtype and has both its ends connected to the reference ground plane. Thefirst notch section 71 consists of a series resonant circuit comprisingan inductance 75, a capacitance 76 and a varactor diode 77. The anodeend of the varactor diode 77 is connected to the reference ground planethrough a series capacitance 78 (RF bypass).

The second notch section 72 consists of a series resonant circuitcomprising an inductance 79, a capacitance 80 and a varactor diode 81.The cathode end of the varactor diode is connected directly to thereference ground plane.

A first tuning voltage V_(T1) is applied to the anode end of thevaractor diode 77 within the first notch section 71 throug a seriesconnection of an RF-bypass capacitance 82 and a limiting resistance 83.The function and operation of the varactor diodes 77 and 81 are similarto those described for the diodes 57 and 61 in FIG. 8. The first tuningvoltage V_(T1) has a fixed value and reverse-biases the varactor diode77 by a fixed amount. As a result, the diode 77 presents a certain valueof junction capacitance into the resonant circuit of the first notchsection 71. The value of the voltage V_(T1) is chosen in such a way thatthe resonant circuit resonates at a center frequency corresponding tothe intermediate frequency (IF) of the receiver in which the filter isused.

In FIG. 9, a second tuning voltage V_(T2) is applied to (i) the cathodeend of the varactor diode 77 in the first notch section 71 through an RFchoke 84, and (ii) the anode end of the varactor diode 81 in the firstnotch section 72 through an RF choke 86. The common supply point ofvoltage V_(T2) is also connected to the reference ground plane through ad-c blocking capacitor 85.

As in the operation of the embodiment of FIG. 8, the second tuningvoltage V_(T2) serves to change the existing reverse-bias on thevaractor diodes 77 and 81, respectively, and hence the junctioncapacitances presented to the resonant circuits of the notch sections 71and 72, respectively.

FIG. 10 illustrates graphically several exemplary idealizedcharacteristic responses of the VNR filters shown in FIGS. 8 and 9 as afunction of variation in the second tuning voltage V_(T2). Three filterresponses 90, 91 and 92 are shown for different values of the tuningvoltage V_(T2). The center frequency of the filter is set by a fixedvalue of the first tuning voltage V_(T1). The second tuning voltageV_(T2) is then adjusted to vary the bandwidth or the notch position N₁and N₂.

Because of its simple design, which allows the null points or thenotches to be easily shifted by varying the second tuning voltage, thefilter can be easily adjusted, via controls which can be provided on thefront panel of the receiver, to produce the optimum video picture. Bymaking these adjustments, the bandwidth of the VNR filter may be limitedto some extent, but any reduction in the carrier signal modulation ismore than compensated for by the rejection of TI and/or otherinterfering noise. Such adjustments can be made for each channelreceived in a particular area.

Referring next to FIG. 11, there is shown a graphical representation oftypical signal-to-noise characteristic for the TVRO receiver systemshown in FIG. 1, which illustrates how the VNR filter described above isso effective in improving the quality of received video images. As shownin FIG. 11, the signal-to-noise characteristic curve 100 maintains aconstant slope until a threshold level 101 is reached. Below thisthreshold level, the output signal-to-noise ratio S_(o) /N_(o) variesessentially linearly with the input signal-to-noise ratio S_(i) /N_(i).Above the threshold level 101, however, the curve 100 has asignificantly increased slope, and it is within this region that theTVRO receiver generally operates.

In this region of increased slope, even a small increase Δi in the inputsignal-to-noise ratio produces a larger change Δ_(o) in the outputsignal-to-noise ratio. Consequently, by suppressing a portion of thenoise at the receiver input the filter according to the presentinvention brings about a dramatic improvement in the outputsignal-to-noise ratio, thereby producing a significantly improved videosignal.

Referring next to FIG. 12, there is shown a switchable version of a VNRfilter according to the system of this invention. This switchable filter110 essentially comprises a first notch section 111 and a second notchsection 112 connected in parallel fashion across a length ofconventional 50-ohm-type strip transmission line 113 and a referenceground plane 114. The first notch section 111 consists of a seriesresonant circuit comprising an inductance 115 and a serially connectedpair of capacitances 116 and 117. The junction of the capacitances isconnected to the reference ground plane through a conventional PIN diode118 with the anode end of the diode toward the ground plane. The secondnotch section 112 consists of a series resonant circuit comprising aninductance 119 and a serially connected pair of capacitances 120 and121. The junction of the capacitances 120, 121 is connected to thereference ground plane through a conventional PIN diode 122 with theanode end of the diode toward the ground plane. As is well known, thePIN diodes act as ordinary diodes at frequencies up to about 100 MHz.However, at the microwave frequencies used for the transmission of TVsignals, which is the applicable frequency range for the filter of thisinvention, the PIN diodescease to act as rectifiers and instead behavelike variable resistances.

When the bias on such a PIN diode is varied, its microwave resistancechanges from a typical value of 5 to 10K ohms under negative bias, tothe vicinity of 1 to 10 ohms when the bias is positive. This variableresistance property of a PIN diode can be used to make it perform likean electronically controlled switch.

Returning to FIG. 12, a control voltage V_(c) is applied through alimiting resistance 123 and a blocking capacitor 124 to (i) the cathodeend of the PIN diode 118 within the first notch section 111 through anRF choke 125, and (ii) the cathode end of the PIN diode 122 within thesecond notch section 112 through an RF choke 126.

When the control voltage V_(c) is low (0 volts), the PIN diodes areeffectively reverse biased and hence present a very high resistance. Inthis mode, the resonant circuits of the two notch sections 111 and 112are controlled by the cumulative effect of the series capacitors 116,117 and 120, 121, respectively.

When the control voltage V_(c) is high (typically 5 volts), the bias onthe PIN diodes 118 and 122 is shifted into the forward region and theresistance of the diodes is switched from the preceding high value to anextremely low value. This effectively shorts the capacitance 117 in thefirst notch section and the capacitance 121 in the second notch sectionout of their respective resonant circuits. The resulting change in theoverall capacitances of the resonant circuits shifts their effectiveresonant frequencies. This produces a corresponding change in the notchfrequencies of the notch sections and thus the overall response of thefilter 110 is shifted.

What is claimed is:
 1. A TVRO receiver for receiving frequency modulatedvideo signals centered within a frequency range having a nominal centerfrequency, the receiver comprising:a tuner including a superheterodynecircuit having a voltage-controlled oscillator (VCO) , means forsupplying a controlling input voltage to said VCO , and a mixer forcombining incoming 1st IF signals within a predetermined IF frequencyrange with the output of said VCO to reduce the frequency of the 1st IFsignal to within a 2nd IF frequency range having a predetermined nominalcenter frequency which permits the output frequencies of said VCO to besubstantially non-interfering with the frequency range of the 1st IFsignals, thereby preventing the output of the VCO from interfering withthe 1st IF signals, and linear phase passband filter means for passing asingle video channel in the second IF input from said mixer, said filtermeans having characteristics with a peak at the center of the filterpassband and having sharply sloping passband edges on either sidethereof, so as the essentialy pass only those signals in the 2nd IFoutput from said mixer which are centered about the center of saidfilter passband.
 2. The TVRO receiver of claim 1 wherein said linearphase passband filter means has passband characteristics which producean output which is at least about 10 dB down from its peak at about +10MHz and -10 MHz from the center of the passband of said filter means. 3.A TVRO receiver for receiving frequency-modulated video signals centeredwithin a frequency range having a nominal center frequency, the receivercomprising:a tuner including a superheterodyne circuit having avoltage-controlled oscillator (VCO), means for supplying a controllinginput voltage to said VCO to produce a corresponding output signal, anda mixer for combining incoming 1st IF signals within a predetermined IFfrequency range with the output signals of said VCO to reduce thefrequency of the 1st IF signals to within a 2nd IF frequency range, thecenter frequency of said 2nd IF frequency range being selected to besuch that the output signals of said VCO are substantiallynon-interfering with the 1st IF signals, and linear phase passbandfilter means for passing a single video channel in the second IF inputfrom said mixer, said filter means having passband characteristics ofthe "haystack" type thereby producing an output which has a peak valuecentered about the passband center, said output being sharply reducedfrom the peak value on either side of the passband center.
 4. The TVROreceiver of claim 3 wherein said linear phase passband filter means haspassband characteristics which produce an output which is at least about10 dB down from its peak at about +10 MHz and -10 MHz from the center ofthe passband of said filter means.
 5. A TVRO receiver for receivingfrequency-modulated video signals centered within a frequency rangehaving a nominal center frequency, said frequency range also includingfrequencies toward one side of the range where terrestrial interference(TI) signals are generally located, the receiver comprising:a tunerincluding a superheterodyne circuit having a voltage-controlledoscillator (VCO), means for supplying a controlling input voltage tosaid VCO, and a mixer for combining incoming 1st IF signals within apredetermined IF frequency range with the output of said VCO to reducethe frequency of the 1st IF signals to within a 2nd IF frequency rangehaving a predetermined nominal center frequency, and linear phasepassband filter means for processing the 2nd IF frequency outputproduced by said mixer, said filter means having passbandcharacteristics with a peak at the center of the filter passband andhaving sharply sloping passband edges on either side thereof, so as toessentially pass only those signals in the 2nd IF output from said mixerwhich are centered about the center of said filter passband therebyeffectively rejecting said TI signals located on one side of thefrequency range of said received signals.
 6. The TVRO receiver of claim5 wherein said linear phase passband filter means has passbandcharacteristics which produce an output which is at least about 10 dBdown from its peak at about +10 MHz and -10 MHz from the center of thepassband of said filtermeans.
 7. A TVRO receiver for receivingfrequency-modulated video signals centered within a frequency rangehaving a nominal center frequency, said frequency range also includingfrequencies toward one side of the range where terrestrial interference(TI) signals are generally located, the receiver comprising:a tunerincluding a superheterodyne circuit having a voltage-controlleroscillator (VCO), means for supplying a controlling input voltage tosaid VCP to produce a corresponding output signal, and a mixer forcombining incoming 1st IF signals within a predetermined IF frequencyrange with the output signals of said VCO to reduce the frequency of the1st IF signals to within a 2nd IF frequency range, and linear phasepassband filter means for processing the 2nd IF frequency outputproduced by said mixer in such a way that only said receiverfrequency-modulated video signals are passed while said TI signalslocated on one side of the frequency range of received signals arerejected, said filter means having passband characteristics of the"haystack" type thereby producing an output which has a peak valuecentered about the passband center, said output being sharply reducedfrom the peak value on either side of the passband center.
 8. The TVROreceiver of claim 2 wherein said linear phase passband filter means haspassband characteristics which produce an output which is at least about10 dB down from its peak at about +10 MHz and -10 MHz from the center ofthe passband of said filter means.